It is a common practice to use transformers to provide power for electronic circuits from the mains supply. This method provides isolation, good regulation, protection from sudden mains fluctuations, and small harmonic distortion in the voltage waveform associated with the waveform of current drawn from the supply. The disadvantages of using transformers in high power application is reflected in their size, weight and winding losses, and for these reasons users often elect to operate directly from the mains supply without employing a transformer.

Where the peak mains potential is adequate for the required application, it is not unusual to obtain a D.C. supply by full wave rectification of the mains with a filter capacitor connected across the rectified supply to reduce ripple. With this arrangement the capacitor charges only when the peak mains voltage exceeds the capacitor voltage, resulting in a large current surge into the capacitor at each peak of the mains voltage. The resulting mains current is a series of pulses separated by equal intervals, and these sudden surges in current tend to distort the sinusoidal shape of the mains voltage, increasing the harmonic content of the supply, and resulting in a poor power-factor. A large magnetic choke placed before the full wave bridge can be used to filter out the higher harmonics of the current pulses while passing the fundamental component, however, such a choke introduces losses and is bulky. A more practical approach is to use the full portion of

the fully rectified mains voltage, for charging the capacitor. This can be achieved by using a switching regulator, in which case the current is distributed over a full cycle and is sinusoidal, and with this type of circuit it is possible to produce an output voltage which is higher than the peak input voltage.

The circuit of the present invention combines the function of a switching regulator with a half-bridge inverter to form a solid state ballast for fluorescent and HID lamps. The disadvantage of each of the circuits described above is that the voltage across the load is essentially D.C, whereas the efficiency of fluorescent and gas discharge lamps increase with higher supply voltage frequencies and therefore an advantage can be gained by using an inverter to drive such lamps.

According to a first aspect, the present invention consists in a power converter, including rectification means to convert an alternating supply potential into a rectified supply potential, and inverter means connected across said rectified supply potential, said inverter means comprising an a.c. divider network having an output which substantially remains at a potential proportional to the rectified supply potential, a half-bridge switching circuit having an output from which alternating potential is produced, said alternating potential having a frequency.substantially higher than that of the supply potential, said switching circuit being coupled across the rectified supply potential at the frequency of said alternating potential but substantially isolated from the rectified supply potential at the frequency of the supply potential, storage means being connected across the switching circuit to maintain a substantially d.c. potential across the switching circuit thereby ensuring that said alternating potential is substantially constant in amplitude, the switching circuit output and the a.c. divider output defining respective sides of the output of said

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inverter means.

The present invention also provides ,a solid state ballast which incorporates the power converter defined above. According to a second aspect, the present invention consists in a switching regulator comprising rectification means to convert an alternating potential of an electrical supply into a rectified supply potential, an inductor and switching element connected in series across the rectified potential, a diode, the anode of which is connected to the junction of the inductor and the switching element and the cathode of which defines the output of the regulator, and storage means being connected across the regulator output, the switching element being controlled by a pulsed switching signal provided by a switching control circuit, the pulsed signal having a frequency which is controlled to increase with decreasing voltage at the regulator output, said regulator being characterised in that a parameter of the pulse signal is varied in response to the instantaneous rectified supply potential to control the waveform of the current flowing from the supply.

In one embodiment of the invention the pulse width is controlled to be inversely proportional to the input voltage and the frequency is proportional to the error in the output voltage, while in another embodiment the pulse width is constant and the frequency is proportional to both the input voltage and the output error voltage.

Embodiments of the invention will now be described by way of example with reference to the accompanying drawings in which: Figure 1 illustrates the circuit schematic of a first embodiment of the invention;

Figure 2 graphically illustrates the current drawn by the circuit of Figure 1 when capacitor C ₃ , is chosen to be too large; Figure 3 graphically illustrates the current drawn by

the circuit of Figure 1 when capacitor C ₃ , is correctly chosen;

Figure 4 schematically illustrates a switching regulator according to a second form of the invention; Figure 5 illustrates several waveforms representing signal levels in the circuit of Figure 3 for (a) high and (b) reduced load conditions when both the pulse width and frequency are controlled;

Figure 6 illustrates similar waveforms to those of Figure 5 (a) and (b) for a system where only the pulse frequency is controlled;

Figure 7 schematically illustrates the regulator of Figure 4 in greater detail; and

Figure 8 schematically illustrates another embodiment of a regulator made in accordance with the present invention. Referring to Fig. 1, a solid state ballast of the present invention includes a bridge rectifier D1-D4 connected to the mains supply to produce a full-wave rectified voltage waveform between the points B and C. A capacitor C,, is connected between the points B and C via a pair of diodes Dr and Dg, such that the capacitor C ₃ , is isolated from the points B and C except when the voltage _BC across these points exceeds the voltage across C ₃ . The voltage V-.„ across C ₃ , is substantially constant with a small ripple due to the discharging of the capacitor C3 between the peaks in V„ _c and the recharging of the capacitor C ₃ when V_ _c approaches its peak value.

A half bridge inverter circuit is connected between the points B, C, D and F and comprises a pair of capactiors C, and C ₂ which form an AC voltage divider, and a pair of transistor switches Q-^ and Q ₂ which alternately switch on to apply either the potential at point D or that at point F to point A. The AC divider is arranged to produce a voltage „ _c between points E and C which is substantially equal to 1/2 V _βC , and as a result, the voltage V _AE which forms the

output of the inverter is a square wave having a peak to peak voltage swing equal to V _Dp and the average value of said square wave being modulated by the voltage - V„ _c .

Each of capacitors C . and C ₅ , which are connected respectively across diodes D _ς and D _g provides a high frequencybypass around its respective diode such that points D and F are isolated from points B and C by the diodes D ₅ and Dg at the ripple frequency of V _BC but are connected via C . and C ₅ at the switching frequency of the transistors Q, and Q _2< In the preferred embodiment, the switching frequency is 25 KHz, however, the value of this frequency is not essential to the operation of the circuit.

The inverter circuit also includes diodes D-, and D« which prevent the voltage V„ _E between points B and E and the voltage „ _c between points E and C from becoming negative in value by more than one diode voltage drop, while diodes D„ and D, _Q serve to protect Q, and Q ₂ from voltage polarity reversals between points D and A, and A and F respectively. In operation the inverter output current flowing between points A and E is drawn, predominantly, from the mains via C ₄ and C _ς each of which have a low impedance at the inverter switching frequency. Capacitor C ₃ , which is an electrolytic capacitor, serves to maintain a substantially constant potential between the points D and F. The values of capacitors C ₃ , C. and C ₅ are chosen to suit the output load connected to the circuit and are selected such that the current drawn from the supply approaches a sinosoidal waveform. Referring to Fig. 2, when C ₃ is chosen to be too large the current drawn from the mains supply will have a sharp peak 10 during the period when capacitor C ₃ is charging. When C ₃ is chosen to be too large, the fall in capacitor voltage due to discharge through the inverter load is so small that the diodes D ₅ and D _g are only forward biased

for a brief period at each peak of the full-wave rectified voltage V _βC . As a result, the capacitor C ₃ must fully charge during this brief period, and as a larger capacitor will have a lower impedance the capacitor will be capable of charging at a sufficiently high rate to create a large peak in the current drawn from the supply.

During the period when C ₃ is not charging, inverter output current is drawn from both the mains supply and from C ₃ such that the current drawn from the mains has a substantially sinusoidal waveform 11 and, referring to Fig. 3, when C ₃ is correctly chosen the capacitor charging current 10 will not seriously affect the sinusoidal shape of the current waveform.

When C ₃ is chosen to be too small the voltage V_ _p will have excessive ripple, resulting in unacceptable variations in the peak to peak value of the square wave component of inverter output voltage V, _E which in turn cause flicker in the light output at the lamp.

Returning to Fig. 1, the inverter load comprises inductor L*., capactior Cg and a fluorescent lamp P. Capacitor C _g and inductor L,, have values which are chosen to allow the combination to resonate at the inverter switching frequency. During the period before the lamp strikes, current flows through the capacitor C _g via both heater elements of the lamp and the voltage across C _g will reach a peak value which allows the tube to strike. After the striking of the tube, the capacitor Cg is bypassed by the tube and the inductor -, , serves to limit the current through the tube. When the circuit of the present invention is employed to drive fluorescent or gas discharge lamps, dimming of the lamp is readily achieved by reducing the supply voltage to the rectifier D,-D..

Figure 4 illustrates a circuit operated directly from the mains to produce a constant output d.c. voltage, and

which can be used to supply the inverter for fluorescent lamps. The circuit of Figure 4 draws a substantially sinusoidal current from the mains supply.

With reference to Figure 4, the mains voltage is fully rectified via diode bridge D21-24. A switching regulator current control (SRCC) circuit produces pulses with period inversely proportional to the amplitudes of the full wave voltage and the frequency proportional to the output current drawn by the load 26. Transistor Q21 is driven as a switch via the oscillator (SRCC and resistor R ₂₁ ) so that when it is "ON", diode D25 is reverse biased, and current in the inductor LlO rises linearly storing energy that is equal to 1/2 LI where L = inductance of ₂₁ I = peak current in the inductor L ₂₁

As Q ₂₁ switches "off", this stored energy is released via diode D _2ς charging capacitor C ₂ -,.

A series of such pulses will maintain the capacitor charge, hence the load voltage across the capacitor to a value that is above the peak full wave voltage V _p , hence diode D ₂₅ is always non-conductive when Q ₂₁ is in the "ON" state.

The driving oscillator circuit (SRCC) monitors the D.C. voltage across the load via feedback line M, and adjusts its frequency accordingly to regulate the voltage, according to the load requirements. Delivered output is proportional to the regulator frequency. Stored energy in coil L ₂ , is proportional to the duration of the driver pulse and the amplitude of the applied voltage. Since amplitude of the input voltage varies as a full-wave, the switching pulse width is chosen to be larger during the initial portion of the full-wave and is reduced with increasing input voltage, throughout the full cycle, see (Fig. 5.). It should be noted that changing the number of pulse trains within a cycle does not affect the relationship of pulse width to the amplitude

of the input voltage.

An alternative choice in switching may be adopted whereby the pulse width is kept constant and only the pulse frequency varied, the number of pulses delivered during the initial portion of the full wave being large, but decreasing in inverse porportion to the amplitude of the full wave. This is illustrated in Fig. 6. The power delivered to the load can be altered by altering the total number of pulses within each cycle. Referring now to Fig. 7, the design of a pulse width modulated switching regulator will be discussed in greater detail. IC, is used as an astable multi vibrator where the duration of each output pulse is determined by the time taken for capacitor C ₃₃ to charge, this charging time being controlled by the series resistor combination 3, R and transistor Q ₃ which is driven directly from the fully rectified wave at node G, via resistor R3g- To correct for the phase shifting effect of the signal at node J, due to base-emitter junction capacitance of Q33, a series network consisting of R37, C3J, is put in parallel with R33.

Thus, as the voltage at node K approaches the 2/3 Vcc value, the voltage at node L stays near Vcc. When the threshold voltage is reached at node K, it is detected by the internal comparator of the 555 I.e., which in turn forces voltages on pins 3 and 7 to zero. Diode D ₃₇ , prevents capacitor C33 from discharging through pin 7, while currents through resistors ₄ 3 and ^ ₄ are returned to the supply ground via the pin 7. Transistors Q3, and Q ₃₂ determine discharge rate of the capacitor C33, which determines the time interval between each of the output pulses on pin 3. Two feed back networks monitor the d.c. - voltage at node M.

One such network consists of R.,, R ₃₃ and D ₃₅ providing a "fast" response, while the other consists of R32' ^R 34' ^R 35 ^an ^ ^c 36 P ^rov •■■*•--•- ng ^a "slow" response.

Slow variations of the d.c. voltage on node M are monitored by the latter network consisting of C3 , R ₃₂ R 4 and R ₃ _, so that an increase in V, effectively drives Q3, further into the "ON" state. At the same time, transistor Q ₃₂ is driven into the "off" state, increasing the discharge rate of capacitor C33, which reduces the frequency of the inverter, and V. is reduced accordingly to a steady value.

Sudden variations in V, can be expected such as when the load is removed. The network consisting of R31, R33 and D ₃₅ monitors such disturbances and regulates V, accordingly.

R ₄₆ and D ₃ g ensure that Q ₃₂ is "OFF" while signal on pin 3 is "high", thus the charging period of C3 is made totally independent of the discharge period.

Q34, L ₃₁ and D ₃ g step up the full-wave at node G into the required, voltage according to the signal on node L of the system described.

The need for a "slow" feed back network arises from the ripples present on , Such ripples are caused by variations in the intensity of energy delivered to the capacitor C ₃₅ during each cycle, and would be considered as small variations in V, by the feedback sensor transistors, Q ₃₁ and Q _32/ resulting in distortion of the mains current. To avoid this condition capacitor C _3g smoothes out the ripple content of the feed back signal, therefore allowing a uniformly symmetrical mains current to be drawn.

Turning now to Figure 8, the hard wired Switching Regulator Current Control Circuit is replaced by a Microprocessor based SRCC circuit wherein the modulation voltage V„ and the feedback voltage V„ are connected to the inputs-of an analog to digital convertion (ADC) 101 which allows these signals to be monitored by the Micoprocessor (UP) 102. The microprocessor 102 calculates the required pulse width and frequency parameters and uses these to

control the Pulse Width Modulator (PWM) 103 in order to produce the SRCC output signal V _L which drives transistor Q ₅₅ via resistor R ς.

The processor is also provided with a serial I/O 5 communications port Dg which can be used to remotely control the SRCC while various other control voltage inputs and control I/O circuits are provided to enable flexible usage of the regulator.

As the input current of the circuit of Figure 8 is 10. determined by the program contained in the microprocessor 102, this circuit lends itself to applications where "shaping" of the input current is required, under these circumstances the system can be programmed to "consume" only the desired portions of the mains cycle, hence any current 15 shape can be produced over a part or all of the mains cycle. It will be recognised by persons skilled in the art that numerous variations and modifications may be made to the invention as described above without departing from the spirit or scope of the invention as broadly described.